Wednesday, May 2, 2012

JLH Class A Amplifier (1969)

Simple Class A Amplifier

A  10-W  design giving  subjectively  better results  than  class  B
transistor amplifiers

by J. L. Linsley Hood, M.I.E.E.

Wireless World, April 1969

During  the past  few  years  a number  of  excellent  designs  have been published  for  domestic  audio amplifiers. However, some of these designs are now rendered obsolescent by changes in the availability of components, and others are  intended  to provide  levels of power output which are  in excess of  the requirements of a normal living room. Also, most designs have tended to be rather complex. In  the circumstances  it seemed worth while  to consider  just how simple a design could be made which would give adequate output power together with a standard of performance which was beyond reproach, and this study has resulted in the present design.

Output power and distortion 

In view of the enormous popularity of the Mullard "5-10" valve amplifier, it appeared that a 10-watt output would be adequate for normal use; indeed when two such amplifiers are used as a stereo pair, the total sound output at full power can be quite astonishing using reasonably sensitive speakers.

The original harmonic distortion standards for audio amplifiers were laid down by D. T. N. Williamson in a series of articles published in Wireless World in 1947 and 1949; and the standard, proposed by him, for less  than 0.1%  total harmonic distortion at  full rated power output, has been generally accepted as  the target figure for high-quality audio power amplifiers. Since the main problem in the design of valve audio amplifiers lies in the difficulty in obtaining adequate performance from the output transformer, and since modern  transistor circuit techniques allow  the design of power amplifiers without output transformers, it seemed  feasible  to aim at a somewhat higher standard, 0.05%  total harmonic distortion at  full output power  over  the  range 30Hz-20kHz. This  also  implies  that the output  power will  be  constant over  this frequency range.

Circuit design

The  first amplifier circuit of which  the author  is aware,  in which a  transformerless  transistor design was used  to  give a  standard of  performance approaching  that  of  the  "Williamson"  amplifier,  was  that published in Wireless World in 1961 by Tobey and Dinsdale. This employed a class B output stage, with series  connected  transistors  in  quasi-complementary  symmetry.  Subsequent  high-quality  transistor power amplifiers have largely tended to follow the design principles outlined in this article.

The major advantage of amplifiers of this type is that the normal static power dissipation is very low, and the overall  power-conversion efficiency  is  high.  Unfortunately  there are also  some  inherent disadvantages due to the intrinsic dissimilarity in the response of the two halves of the push-pull pair (if complementary transistors  are used  in unsymmetrical  circuit  arrangement)  together with  some  cross-over distortion due to the Ic/Vb characteristics. Much has been done, particularly by Bailey(1), to minimise the latter.

An additional  characteristic  of  the  class  B  output  stage  is  that the  current  demand of  the output transistors  increases with  the output  signal,  and  this may  reduce  the output  voltage and worsen  the smoothing of the power supply, unless this is well designed. Also, because of the increase in current with output  power,  it  is  possible  for  a  transient  overload  to drive  the output transistors  into a  condition of thermal runaway, particularly with reactive loads, unless suitable protective circuitry is employed. These requirements have combined to increase the complexity of the circuit arrangement, and a well designed low-distortion class B power amplifier is no longer a simple or inexpensive thing to construct.

An alternative approach  to  the design of a  transistor power amplifier combining good performance with simple construction is to use the output transistors in a class A configuration. This avoids the problems of asymmetry  in  quasi-complementary  circuitry, thermal  runaway  on  transient  overload,  cross-over distortion and signal-dependent variations in power supply current demand. It is, however, less efficient than a class B circuit, and the output transistors must be mounted on large heat sinks.

The basic class A construction consists of a single transistor with a suitable collector load. The use of a resistor, as in Fig. 1(a), would be a practical solution, but the best power-conversion efficiency would be about  12%.  An  l.f.  choke,  as shown  in  Fig.  1(b),  would  give  much better  efficiency,  but  a properly designed  component  would be  bulky  and expensive,  and  remove  many  of  the advantages  of  a transformerless design. The use of a second, similar, transistor as a collector load, as shown in Fig. 1(c), would be more convenient in terms of size and cost, and would allow the load to be driven effectively in push-pull  if  the  inputs  of  the  two  transistors were of  suitable magnitude and opposite  in phase.  This requirement can be achieved if the driver transistor is connected as shown in Fig. 2.

This  method of  connection also  meets  one of  the  most  important  requirements  of  a  low  distortion amplifier  -  that the basic  linearity  of  the amplifier  should be  good,  even  in  the absence of feedback. Several factors contribute to this. There is the tendency of the Ic/Vb non-linearity of the characteristics of  the output transistors  to  cancel,  because during  the part  of  the  cycle  in  which one  transistor  is approaching cut-off the other is turned full on. There is a measure of internal feedback around the loop Tr1, Tr2, Tr3 because of the effect which the base impedance characteristics of Tr1 have on the output current  of Tr3. Also, the driver  transistor Tr3, which has  to deliver a  large  voltage  swing,  is operated under conditions  which  favour  low  harmonic  distortion  -  low  output  load  impedance,  high  input impedance.  A practical power amplifier circuit using this type of output stage is shown in Fig. 3.

The open  loop gain of  the circuit  is approximately 600 with  typical  transistors. The closed  loop gain  is determined, at frequencies high enough for the impedance of C3 to be small in comparison to R4, by the ratio (R3 + R4)/R4. With the values indicated in Fig. 3, this is 13. This gives a feedback factor of some 34dB, and an output impedance of about 160 milliohms. 

Since  the circuit has unity gain at d.c., because of  the  inclusion of C3  in  the  feedback  loop, the output voltage, Ve,  is held at the same potential as  the base of Tr4 plus  the base emitter potential of Tr4 and the small potential drop along R3 due to the emitter current of this transistor. Since the output transistor Tr1 will turn on as much current as  is necessary  to pull Ve down  to  this value, the  resistor R2, which together with R1 controls the collector current of Tr2, can be used to set the static current of the amplifier output stages. It will also be apparent that Ve can be set to any desired value by small adjustments to R5 or R6. The optimum performance will be obtained when  this  is equal  to half  the supply voltage. (Half a volt or so either way will make only a small difference to the maximum output power obtainable, and to the other characteristics of the amplifier, so there is no need for great precision in setting this.) 

Silicon planar transistors are used throughout, and this gives good thermal stability and a low noise level. Also, since  there  is no  requirement  for complementary symmetry, all  the power stages can use n-p-n transistors which offer,  in silicon, the best performance and  lowest cost. The overall performance at an output level of 10 watts, or at any lower level, more than meets the standards laid down by Williamson. The power  output  and  gain/frequency  graphs  are  shown  in Figs.  4  -  6,  and  the  relationship between output power and  total harmonic distortion  is shown  in Fig. 7. Since  the amplifier  is a straight-forward class A circuit, the distortion decreases  linearly with output voltage. (This would not necessarily be  the case in a class B system if any significant amount of cross-over distortion was present.) The analysis of distortion components at levels of the order of 0.05% is difficult, but it appears that the residual distortion below the level at which clipping begins is predominantly second harmonic

Stability, power output and load impedance 

Silicon planar  n-p-n  transistors  have,  in  general,  excellent  high  frequency  characteristics,  and  these contribute  to  the very good stability of  the amplifier with reactive  loads. The author has not yet  found a combination of  L and C which makes  the  system  unstable,  although  the  system will  readily  become oscillatory  with an  inductive  load  if  R3  is shunted by  a  small  condenser  to  cause  roll-off  at  high frequencies.

The circuit shown  in Fig. 3 may be used, with very  little modification  to  the component values, to drive load  impedances  in  the  range 3  -  15 ohms. However, the  chosen output  power  is  represented by  a different current/voltage relationship in each case, and the current through the output transistors and the output-voltage swing will  therefore also be different. The peak-voltage swing and mean output current can be calculated quite simply from the well-known relationships W=I2.R and V=I.R, where the symbols have their  customary  significance.  (It  should be  remembered,  however, that the  calculation of  output power is based on  r.m.s. values of current and voltage, and  that these must be multiplied by 1.414  to obtain the peak values, and that the voltage swing measured is the peak-to-peak voltage, which is twice the peak value.) 

When these calculations have been made, the peak-to-peak voltage swing for 10 watts power into a 15- Ohm load is found to be 34.8 volts. Since the two output transistors bottom at about 0.6 volts each, the power  supply must provide a minimum of 36  volts  in order  to allow  this output. For  loads of 8 and 3 ohms, the minimum h.t.  line  voltage must be 27V and 17  volts  respectively. The necessary minimum currents  are 0.9,  1.2 and  2.0 amps.  Suggested  component  values  for  operation  with  these  load impedances are shown  in Table 1. C3 and C1  together  influence  the voltage and power roll-off at  low audio  frequencies. These  can be  increased  in  value  if  a better  low-frequency performance  is desired than that shown in Figs. 4 - 6.

Since  the supply voltages and output currents  involved  lead  to dissipations  in  the order of 17 watts  in each output transistor,  and  since  it  is  undesirable  (for  component  longevity)  to permit  high operating temperatures,  adequate heat  sink  area  must  be provided  for  each  transistor.  A  pair  of  separately mounted 5in by 4in finned heatsinks is suggested. This is, unfortunately, the penalty which must be paid for class A operation. For supplies above 30V Tr1 and Tr2 should be MJ481s and Tr3 a 2N1613.

If  the output  impedance of  the pre-amplifier  is more  than a  few  thousand ohms, the  input stage of  the amplifier should be modified  to  include a simple  f.e.t. source  follower circuit, as shown  in Fig. 8. This increases  the harmonic  distortion  to about  0.12%,  and  is  therefore  (theoretically)  a  less  attractive olution  than a better  pre-amplifier.  A  high  frequency  roll-off  can  then be obtained,  if  necessary,  by connecting a small capacitor between the gate of the f.e.t. and the negative (earthy) line.

Suitable transistors

Some experiments were made  to determine  the extent to which the circuit performance was influenced by the  type and current gain of  the  transistors used. As expected  the best performance was obtained when high-gain  transistors were used, and when  the output stage used a matched pair. No adequate substitute is known for the 2N697 / 2N1613 type used in the driver stage, but examples of this transistor type from three different manufacturers were used with apparently identical results. Similarly, the use of alternative  types  of input transistor  produced no apparent  performance  change,  and  the  Texas Instruments 2N4058 is fully interchangeable with the Motorola 2N3906 used in the prototype.

The most noteworthy performance changes were  found in the current gain characteristics of the output transistor pair, and for the lowest possible distortion with any pair, the voltage at the point from which the loudspeaker  is  fed should be adjusted so that it is within 0.25 volt of half the supply line potential.  The other results are summarized in Table 2.

The transistors used in these experiments were Motorola MJ480 / 481, with the exception of (6), in which Texas 2S034 devices were  tried. The main conclusion which can be drawn  from  this  is  that the type of transistor used may not be very  important, but that  if  there are differences  in  the current gains of  the output transistors, it is necessary that the device with the higher gain shall be used in the position of Tr1. When distortion  components were  found prior  to  the onset  of waveform  clipping, these were almost wholly due to the presence of second harmonics.

Constructional notes 

Amplifier. The components necessary  for a 10 + 10 watt stereo amplifier pair can be conveniently be assembled on a standard  “Lektrokit” 4in x 4.75in s.r.b.p. pin board, as shown  in  the photographs, with the  four  power  transistors  mounted on external  heat  sinks.  Except  where noted  the  values  of components do not appear  to be particularly critical, and 10% tolerance resistors can certainly be used without  ill effect. The  lowest noise  levels will however be obtained with good quality components, and with carbon-film, or metal-oxide, resistors.

Power Supply. A suggested form of power supply unit is shown in Fig. 9(a). Since the current demand of  the amplifier  is substantially constant, a series  transistor smoothing circuit can be used  in which  the power supply output voltage may be adjusted by choice of the base current input provided by the emitter follower Tr2 and the potentiometer VR1. With the values of the reservoir capacitor shown in Table 3, the ripple level will be less than 10mV at the rated output current, provided that the current gain of the series transistors is greater than 40. For output currents up to 2.5 amps, the series transistors indicated will be adequate, provided that they are mounted on heat sinks appropriate to their loading.

However, at the current levels necessary for operation of the 3-ohm version of the amplifier as a stereo pair, a single MJ480 will no  longer be adequate, and either a more suitable series  transistor must be used,  such as  the Mullard  BDY20,  with  for  example a 2N1711 as  Tr2,  or  with a parallel  connected arrangement as shown in Fig. 9(b).

The  total  resistance  in  the  rectifier  "primary" circuit,  including  the  transformer secondary winding, must not be  less  than 0.25Ω. When  the power supply, with or without an amplifier,  is  to be used with an r.f. amplifier-tuner  unit,  it may  be necessary  to add  a 0.25uF  (160V.w.)  capacitor  across  the  secondary winding  of  T1  to prevent transient  radiation.  The  rectifier  diodes specified are  International Rectifier potted bridge types.

Transistor protection circuit

The current which  flows  in  the output transistor chain  (Tr1, Tr2)  is determined by  the potential across Tr2, the values of R1 and R2, and the current gain and collector-base leakage current of Tr2. Since both these  transistor  characteristics  are  temperature dependant the output  series current  will increase somewhat with the temperature of Tr2. If the amplifier is to be operated under conditions of high ambient temperature, or  if for some reason  it  is not practicable to provide an adequate area of heat-sink for the output transistors,  it will  be desirable  to provide  some alternative means  for  the  control  of  the output transistor  circuit  current.  This can be done by means  of  the  circuit  shown  in  Fig.  10. In  this,  some proportion of  the d.c.  bias current to  Tr1  is shunted  to  the negative  line  through  Tr7, when  the  total current  flowing causes  the potential applied  to  the base of Tr6  to exceed  the  turn-on value  (about 0.5 volt). This allows very precise control of the series current without affecting the output power or distortion characteristics. The simpler arrangement whereby the current control potential for Tr7 is obtained from a series  resistor  in  the emitter  circuit  of  Tr1  leads,  unfortunately, to a  worsening  of  the distortion characteristics to about 0.15% at 8 watts, rising to about 0.3% at the onset of overload.

Performance under listening conditions

It would be convenient if the performance of an audio amplifier (or loudspeaker or any other similar piece of audio equipment)  could be  completely  specified by  frequency  response and harmonic  distortion characteristics. Unfortunately, it is not possible to simulate under laboratory conditions the complex loads or intricate waveform structures presented  to  the amplifier when a  loudspeaker system  is employed  to reproduce  the everyday  sounds  of  speech and  music;  so  that  although  the  square  wave and  low- distortion sine wave oscillators, the oscilloscope, and the harmonic distortion analyser are valuable tools in the design of audio circuits, the ultimate test of the final design must be the critical judgement of the listener under the most carefully chosen conditions his facilities and environment allow.

The possession of a good standard of reference is a great help in comparative trials of this nature, and the author  has  been  fortunate  in  the possession,  for many years, of a  carefully and expensively built “Williamson” amplifier, the performance of which has proved,  in  listening  trials, to equal or exceed, by greater or lesser margins, that of any other audio amplifier with which the author has been able to make comparisons.

However,  in  the past, when  these  tests were made  for personal curiosity, and some  few minutes could elapse  in  the  transfer  of  input  and output  leads  from  one amplifier  to  the other, the  comparative performance of some designs has been so close  that the conclusion drawn was  that there was  really very little  to  choose between  them.  Some of  the  recent transistor  power  amplifier  circuits  gave a performance which seemed fully equal to that of the “Williamson”, at least so far as one could remember during  the  interval between one  trial and  the next. It was, however, appreciated  that this did not  really offer  the best  conditions  for  a proper  appraisal  of  the more  subtle differences  in  the performance of already good designs, so a changeover switch was arranged to transfer inputs and outputs between any chosen pair of amplifiers, and a  total of six amplifier units was assembled,  including  the  “Williamson”, and another popular valve unit, three class B transistor designs, including one of commercial origin, and the class A  circuit  described above.  The  frequency  response,  and  total  harmonic  distortion characteristics, of the four transistor amplifiers was tested in the laboratory prior to this trial, and all were found to have a flat frequency response through the usable audio spectrum, coupled with low harmonic distortion content (the worst-case figure was 0.15%).

In  view  of  these prior  tests,  it was  not  expected  that there would be any  significant  difference  in  the audible performance of any of  the  transistor designs, or between  them and  the valve amplifiers. It was therefore surprising to discover, in the event, that there were discernable differences between the valve and the three class B transistor units. In fact, the two valve designs and the class A transistor circuit, and the three class B designs  formed  two  tonally distinct groups, with closely similar characteristics within each group.

The “Williamson” and the present class A design were both better than the other valve amplifier, and so close in performance that it was almost impossible to tell which of the two was in use without looking at the switch position. In  the upper reaches of  the  treble spectrum  the  transistor amplifier has perhaps a slight advantage.

The performance differences between  the class A and  the class B groups were, however, much more prominent. Not only did the class A systems have a complete freedom from the slight “edginess” found on some high string notes with all  the class B units, but they appeared also  to give a  fuller, “rounder”, quality, the attractiveness  of which  to  the author much outweighs  the  incidental inconvenience of  the need for more substantial power supply equipment and more massive heat sinks.

Some thought, in discussions with interested friends, has been given to the implications of this unlooked-for discovery,  and a  tentative  theory  has  been evolved  which  is  offered  for  what  it  is  worth. It  is postulated  that these  tonal  differences  arise because  the normal  moving-coil loudspeaker,  in  its associated housing, can present a very complex reactive  load at  frequencies associated with structural resonances, and that this might provoke transient overshoot when used with a class B amplifier, when a point of  inflection  in  the applied waveform chanced  to coincide with  the point of  transistor crossover, at which point,  because of  the abrupt  change  in  the  input  parameters  of  the output transistors  the  loop stability margins and output damping will be  less good. In  these circumstances, the desired  function of the power-amplifier output circuit in damping out the cone-response irregularities of the speaker may be performed worse at the very places in the loudspeaker frequency-response curve where the damping is most needed.

It should be emphasized that the differences observed in these experiments are small, and unlikely to be noticed except in direct side-by-side comparison. The perfectionist may, however, prefer class A to class B in transistor circuitry if he can get adequate power for his needs that way.

Listener fatigue

In  the experience of  the author, the performance of most well-designed audio power amplifiers is really very good, and the differences between one design and another are likely to be small in comparison with the differences between alternative loudspeaker systems, for example, and of the transistor designs so far encountered, not one could be considered as unpleasing to the ear. However, with the growing use of  solid-state power amplifiers, puzzling tales of “listener fatigue” have been heard among the cognoscenti, as something  which all  but the  most  expensive  transistor  amplifiers  will  cause  the  listener,  in contradistinction with good valve-operated amplifiers. This seemed to be worth investigation, to discover whether there was any foundation for this allegation.

In practice  it  was  found  that  an amplifier  with an  impeccable performance on paper  could be  quite worrying  to  listen  to under certain conditions. This appears  to arise and be particularly associated with transistor power amplifiers because most of  these are easily able  to deliver  large amounts of power at supersonic  frequencies, which  the  speakers  in a high quality  system will endeavour  to present to  the listener. In  this context  it should be  remembered  that  in an amplifier which has a  flat power response from 30Hz to 180kHz, 90% of this power spectrum will be supersonic.

This unwanted output can arise  in  two ways. It can be because of wide spectrum “white noise”  from a preamplifier with a significant amount of hiss – this can happen if a valve preamplifier is mismatched into the few thousand ohms input impedance of a transistor power amplifier, and will also cause the system performance to be unnaturally lacking in bass. Trouble of this type can also arise if transient instability or high frequency “ringing” occurs, for example when a reactive load is used with a class B amplifier having poor cross-over point stability.


1.   Bailey,  A.R.,  “High-performance  Transistor  Amplifier”, Wireless World, November  1966;  “30-Watt High Fidelity Amplifier”, May 1968 and “Output Transistor Protection in A.F. Amplifiers”, June 1968.

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