Design by T. Giesberts
Source: Elektor Electronics 4/2001
In early 1984, Elektor magazine beat the competition hands-down by publishing the design of a then-revolutionary MOSFET amplifier. Even now, this amplifier enjoys an enthusiastic following. In response to many requests, we have brought the design up to date and given some attention to improved reliability and operating safety. The output power is 90 watts into 8 ohms or 135 watts into 4 ohms, which should leave little to be desired for most users.
From the number of zeros after the decimal point, you can see in a single glance that this is an exemplary set of results. You will not often come across a better set of figures. The distortion is very low, the damping factor is very good and the slew rate can even be said to be remarkably good. As you may expect, we have also measured a number of curves using the Audio Precision analyser in order to complement the performance figures, which always have a somewhat ‘dry’ taste. Figure A shows the harmonic distortion (THD+N) over the range of 20 Hz to 20 kHz with an 8-Ω load, using a measurement bandwidth of 80 kHz. At 1 W the increase in the distortion level at 20 kHz is minimal, but at the 50% power level (40 W is equivalent to 70% of the maximum output amplitude) the effect of the nonlinear input capacitance of the MOSFETs can be recognised. Figure B shows the distortion of a 1-kHz signal into an 8-Ω load as a function of the output level in watts, measured with a bandwidth of 22 kHz. The behaviour of the amplifier is more readily visible with this narrower measurement bandwidth. Up to 10 W, the THD+N is predominantly due to supply ripple and noise. A slight increase in the distortion can be seen above 10 W, but a level of 0.1% is reached only at 90 W. Figure C shows the maximum output power into 4-Ω and 8-Ω loads at a distortion level of 0.1% for frequencies between 20 Hz and 20 kHz (80 kHz measurement bandwidth). Both of these curves can be said to be practically straight. Finally, Figure D shows the results of a Fourier analysis of a 1-kHz signal (1 W into 8 Ω) with the fundamental suppressed. At this power level, the THD is clearly lower than the supply ripple, whose harmonics lie below –100 dB. The 2nd and 3rd harmonics lie at negligibly low levels (–118 dB and –115 dB, respectively).
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Figure 1A - Schematic diagram of Mini Crescendo 1984 Version |
There are surely not very many circuit designs that continue to enjoy such a high level of interest more than ten years after their original appearance, as does the Crescendo power amplifier from 1984. In part, this is due to its completely symmetrical design, which was in fact an unusual feature at that time, but unquestionably it is mainly due to the use of power MOSFETs in the output stage. A lot of people happen to be fervent fans of these devices. Even people who swear by valve amplifiers and are allergic to anything with ‘semiconductor’ in its name often have a weakness for MOSFETs, and are thus prepared to make an exception for them. Sadly enough, most of the problems with the amplifier in question had to do with the MOSFETs. The original types have long since gone obsolete and become unavailable, and suitable replacements are hard to find. However, there were also other difficulties. The stability of the amplifier sometimes gave cause for concern, and users considered the absence of protective circuitry to be a major weakness. Consequently, in honour of our anniversary, we decided to take another look at the original design. Our objective was to update the design of the amplifier in a way that would eliminate the sources of criticism without sacrificing the good characteristics of the original design. This objective has been quite successfully achieved. In addition, we were able to obtain such a generous level of output power using a new pair of MOSFETs that it is not necessary to split the new Crescendo into ‘light’ and ‘heavy’ versions. The same concept Since we have intentionally tried to change the old amplifier design as little as possible, the differences between the schematic diagrams of the old and new versions are minimal. The design still consists of an input stage with dual differential amplifiers and current sources, a cascode driver stage and a MOSFET output stage. That may have been a rather sophisticated design in 1984, but nowadays it would more likely be described as a ‘minimal design’. There’s nothing wrong with this, by the way, since attempting to keep the signal path as short as possible is certainly not a mistaken endeavour in an amplifier design — but we don’t need to dwell on this point. Since the basic concept of the original design has been retained, anyone who compares the schematic diagram of the new version (see Figure 1) with that of the old version (May 1984) will first have to try to find the differences. Of course, there are indeed differences, and it seems like a good idea to list the most important changes before diving into a detailed description of the schematic diagram. The most evident change is naturally the new pair of MOSFETs in the output stage. The Toshiba 2SK1530 and 2SJ201 are readily available, and furthermore they can dissipate so much more power than the original devices that we were able to boost the output power of the old ‘MiniCrescendo’ by a factor of nearly two (90 W into 8 Ω in place of 50 W) using only a single pair of transistors. As a consequence of the increased power level, the bias currents of the various stages must be modified and different transistors must be used in the cascode stage, as will be seen later on. The next change is the addition of the networks R10/C4, R15/C5 and R30/R31, which represent the results of measures that have been taken to optimise the stability of the amplifier. A very important final item is that the amplifier has been provided with reliable protection circuitry and automatic offset compensation, by means of an extra printed circuit board. This pretty well covers the most important changes.
Schematic details
Now that we’ve seen the global picture, it’s time to take a more detailed look at the circuit diagram. Let’s start at the beginning, which is of course the input stage. The design of the input filter is more or less standard. R2 (with R1 in parallel) determines the input impedance, and in combination with C1 it forms a high-pass filter that blocks frequencies below around 1.5 Hz. C1 is also needed to isolate the DC bias of the input stage. The combination of R3 and C2 forms a low-pass filter that is dimensioned for a frequency of more than 300 kHz. This helps prevent TIM (transient intermodulation) distortion and eliminates possible RF interference. The dual differential amplifier (T1–T4) has been designed to work with a bias current that is approximately three times a great as that of the original design, on account of the increased output power. The current sources that regulate this setting, T5 and T6, now use LEDs as references (D1 and D2), since this results in less noise than using Zener diodes. In the interest of the thermal stability of the DC setting, D1/T5 and D2/T6 are thermally coupled, as are the transistor pairs T1/T2 and T3/T4. The bias currents of the cascode stages T7/T8 and T9/T10 are also significantly greater than in the original design. Since this would be a bit too much for the transistor types used for T8 and T10 in the old version, they have been replaced by the somewhat more robust types MJE340 and MJE350. Now we come to the output stage. In contrast to the MOSFETs used in the old version, the 2SK1530 and 2SJ201 devices used here have a positive temperature coefficient. This means that with a constant gate-source voltage, the drain current increases with increasing temperature. This made it necessary to use a different design for the quiescent-current circuit. Here the MOSFET T11, which is mounted on the same heat sink as T8/T10 and T12/T13, provides the necessary compensation. Finally, there are a couple of other significant items. Insiders will notice that the none-too-attractive bipolar electrolytic capacitor has been eliminated from the reverse feedback network (R22/R23), which means that DC coupling is used here. To get rid of the resulting output offset, we have provided an automatic compensation circuit that is located on the protection circuit board.
We anyhow intended touse the compensation circuit to correct for the offset caused by the unavoidable asymmetry of the input stage. The necessary compensation circuit consists of nothing more than an opamp wired as an integrator, which measures the output voltage of the amplifier and provides the proper amount of reverse current feedback to the (bias) input. Thanks to the very high values of R4 and R5 and the decoupling provided by C3, this correction has absolutely no effect on the audio signal. Another essential detail is that the open-loop gain has been made independent of the load by the addition of R30 and R31. These resistors together determine the output impedance of the voltage amplifier, and as a result the source followers T12 and T13 now operate purely as buffers in the audio range. Without these resistors, the behaviour of the amplifier is directly dependent on the connected load, which is not the way things are supposed to be. Together with the compensation networks R10/C4 and R15/C5, the modification made using R30/R31 ensures that the amplifier is unconditionally stable, so much so that the standard Boucherot network (R36/C11) can even be omitted.
Protection

A robust power supply

two of these supplies! The ‘mains switch-on delay’ shown inside the dotted box in Figure 3 is not mandatory, but it is highly recommended — especially if a toroidal transformer is used. This circuit does exactly what its name suggests, and it ensures that excessive current surges do not occur when the mains voltage is switched on. Such circuits have frequently been described in Elektor Electronics; the most recent one can be found in the Summer Circuits issue of 1997, and we have reproduced its schematic diagram in Figure 4. Its operation is simple, and is based on the fact that the current is initially limited by R4-R7 immediately after switch-on. After the expiry of a time delay determined by C2 and C3, these resistors are bridged over by the relay and the full current flows between K1 and K2. The relay used here is a type that can switch 2000 VA. The supply voltage for the relay is taken directly from the mains circuit via C1, R3 and B1, so this circuit is dangerous to the touch!
Soldering

Wiring and set-up


terminal of the protection board to the positive output socket (banana socket). The other (negative) banana socket is connected directly to the ‘LSP–‘ terminal. The necessary connection between the circuit ground of the amplifier and the metallic enclosure can best be realised by fitting the Cinch (a.k.a. RCA or ‘line’) input sockets in a ‘normal’ (non-insulated) manner. Take care that there is not any other unintentional connection between the signal ground and the enclosure ground, since this will create an earth loop that can cause stubborn hum problems. It goes without saying that a well-insulated cable, a robust mains switch and an equally robust mains entrance must be used for the connection to the 230-V mains circuit. Pay attention to the electrical safety of the overall assembly, and attach an identification label that lists the specified values of the supply voltage (230 V) and fuse to the outside of the enclosure. Once you have again thoroughly checked everything and re-measured the supply voltages, it’s nearly time to power up the amplifier. Before doing this, however, you must turn trimpot P1 fully to the left (counter clockwise). Otherwise you run the risk that the quiescent current will immediately rise to a very high level, which is not what we want. After switching on the unit, first check the amplifier output (test point tp3) to verify that the voltage is zero. An offset of a few millivolts is acceptable, but if you measure 0.1 V or more you will have to carefully reinspect the whole assembly, since there is something wrong. Following this, you can set the quiescent current to the proper value. The ideal value for this amplifier is 200 to 250 mA. To adjust the quiescent current, connect a voltmeter across R34 (test points tp1 and tp3) and turn P1 slowly until the measured voltage is between 0.044 and 0.055 V. Then let the amplifier warm up for half an hour, and again adjust the current to the same value using P1.
Listening

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